Transconductor and filter circuit using the same

ABSTRACT

A filter circuit capable of suppressing an increase of circuit area, enabling easy circuit design, realizing a reduction of power consumption by a common control voltage operation, and able to stably control the cut-off frequency, that is, a filter circuit including a Gm-C circuit comprising a plurality of transconductors and load capacitors driven by the transconductors, where control signals (bias voltages) to be supplied to the transconductors are controlled by bias circuits in accordance with output voltages output from the transconductors and a reference voltage input from the outside so that output common-mode potentials of the transconductors become identical, the common-mode input potentials are controlled so that g m  values become desired values in the transconductors, and W/L ratios of MOS transistors of a differential pair are set to be predetermined values.

BACKGROUND OF THE INVENTION

[0001] 1. Field of the Invention

[0002] The present invention relates to a filter circuit, more particularly relates to an active filter circuit called a transconductor-C (Gm-C), and a transconductor serving as a component of a filter circuit.

[0003] 2. Description of the Related Art

[0004] In an integrated filter circuit including an active filter, for example, a Gm-C filter, it is desirable to enable easy, linear adjustment of a cut-off frequency f_(c) while maintaining a Q factor of the filter.

[0005] A waveform equalizing technique as represented by partial response-maximum likelihood (PRML) is generally applied to a stored data reproduction system (read channel) for reproducing stored information from an information storage medium such as a magnetic or optical disk. Normally, in a signal waveform reproduced via an optical pickup or a magnetic head from a storage medium, there is large signal leakage between adjacent bit data, namely, large inter-symbol interface (ISI), so reproduction of data only by the signal level of the sampling time is difficult. The partial response (PR) and other techniques enable high density recording and reproduction combined together with a later stage Viterbi decoding algorithm etc. by permitting ISI only for two to five adjacent sampling times, while eliminating signal leakage in other sampling times.

[0006] By taking as an example a magnetic medium wherein a reproduced signal is inherently a differential system, the equalization method used is a differential series such as PR4 (equalizing a write code 1 to three adjacent sample rows 1, 0, and −1), EPR4 (similarly equalizing it to 1, 1, −1, and −1), and EEPR4 (similarly equalizing it to 1, 2, 0, −2, and −1). Specifically, a high frequency enhanced analog low pass filter is used as an equalizer. For example, a 7-pole 2-zero filter comprising a Gm-C biquadratic filter is proposed by Geert A. De Veirman and Richard G. Yamasaki in “Design of a Bipolar 10-MHZ Programmable Continuous-Time 0.05° Equiripple Linear Phase Filter”, IEEE Journal of Solid-State Circuits, vol. 27, no. 3, March 1992. This filter configuration has a linearity of phase characteristic required by a digital read channel, that is, a good, constant group delay characteristic, and is generally used as an analog equalizing filter.

[0007]FIG. 5 is a block diagram of the configuration of the filter.

[0008] As shown in the figure, the filter comprises cascade-connected biquadratic filters/equalizers 101 (biquad1/equalizer), 102 (Biquad2), and 103 (Biquad3) and a low pass filter (LPF) 104. Note that the biquadratic filter/equalizer 101 has an equalizing function. In the filter configuration shown in FIG. 5, a reproduced signal S_(in) is controlled in gain, then input to the first stage biquadratic filter/equalizer 101 and there adjusted in high frequency boost and equalized. Then, together with the biquadratic filters/equalizers 102 and 103 and the low pass filter 104 connected thereafter, a phase characteristic having a constant group delay is attained. According to Veirman and Yamasaki, the pole frequencies and Q factors of the filter components are as shown in FIG. 6.

[0009] The pole frequencies in FIG. 6 are scaled by the cut-off frequency of the equalizing filter. For example, in a read channel having a data rate of 400 Mbps, the cut-off frequency of the equalizing filter becomes about 100 MHZ. As a result, if the cut-off frequency of the equalizing filter is assumed to be 100 MHZ, from FIG. 6, for example, the pole frequency, that is, the cut-off frequency, of the third stage biquadratic filter 103 becomes 231.74 MHZ. Note that the combinations of the pole frequencies and Q factors in FIG. 6, that is, the pole arrangement, are those of “a linear phase filter having a 0.05° equiripple error” well known in filter design, but the invention is also applicable to other combinations of the pole frequencies and Q factors. The pole arrangement here is just an example.

[0010] The reproduced data rate of a disk medium differs by about 2.5 times between its inner track and outer track and is required to be adjustable to an optimal cut-off frequency by an external control means. At this time, all of the filter components, that is, the biquadratic filters and the low pass filter, have to have Q factors held at the values indicated in FIG. 6 at all times. Further, the ratios of pole frequencies of the biquadratic filters and the low pass filter have to be the ratios indicated in FIG. 6 regardless of the cut-off frequency of the equalizing filter as a whole. In other words, when adjusting the cut-off frequency of the equalizing filter as a whole in accordance with a change of the reproduced data rate, it is necessary that the component biquadratic filters and the low pass filter be monotonously increased or decreased in pole frequencies while maintaining constant Q factors.

[0011] Next, a method of designing the above cut-off frequency and Q factor will be explained by showing an example of the circuits of the components when configuring an equalizing filter by a Gm-C filter.

[0012]FIG. 7 shows the basic configuration of the biquadratic filters 102 and 103, while FIG. 8 shows a feed forward pulse slimming configuration used in the biquadratic filter/equalizer 101. Furthermore, FIG. 9 shows the configuration of a primary low pass filter 104.

[0013]FIG. 7 shows an example of the configuration of a biquadratic filter having a differential configuration. As shown in the figure, two integrators comprised of Gm-C's are connected in cascade, while a negative feedback loop comprised of another Gm cell is connected to an output terminal thereof. Note that in FIG. 7, the load capacitance C is expressed as a differential capacitance, but generally 2C capacitances are connected between positive and negative signal lines and a ground potential for the reason of ease in setting capacitance considering the amount of parasitic capacitance and combining a function of phase compensation capacitance in a common-mode feedback loop.

[0014] The transfer function of the biquadratic filters 102 and 103 having the configuration shown in FIG. 7 and used as an equalizing filter is given by the formula below: $\begin{matrix} {\frac{Vlp}{Vi} = \frac{g_{m1}{g_{m3}/C^{2}}}{s^{2} + {s\left( {g_{m2}/C} \right)} + \left( {g_{m1}{g_{m3}/C^{2}}} \right)}} & (1) \end{matrix}$

[0015] Accordingly, the pole frequency ω₀ and Q (quality factor) of a filter are expressed by the formulas below: $\begin{matrix} {{\omega_{0} = \frac{\sqrt{g_{m1}g_{m3}}}{C}},{Q = \frac{\sqrt{g_{m1}g_{m3}}}{g_{m2}}}} & (2) \end{matrix}$

[0016]FIG. 8 shows an example of the configuration of an equalizing filter comprising an equalizer unit capable of adjusting a high pass boost by a feed forward amplifier K. The transfer function of the equalizing filter is given by the formula below. $\begin{matrix} {\frac{Vlp}{Vi} = \frac{\left( {g_{m1}{g_{m3}/C^{2}}} \right) - K_{s}^{2}}{s^{2} + {s\left( {g_{m2}/C} \right)} + \left( {g_{m1}{g_{m3}/C^{2}}} \right)}} & (3) \end{matrix}$

[0017] Similarly, the pole frequency ω₀ and Q of the filter are expressed by the formulas below: $\begin{matrix} {{\omega_{0} = \frac{\sqrt{g_{m1}g_{m3}}}{C}},{Q = \frac{\sqrt{g_{m1}g_{m3}}}{g_{m2}}}} & (4) \end{matrix}$

[0018] Here, the reason for realizing the high pass boost by the biquadratic filter/equalizer 101 is, as will be understood from FIG. 6, that high pass boosting can be attained by a relatively small K. Therefore, realization of high pass boosting is not limited to the biquadratic filter/equalizer 101 and can be attained by other biquadratic filters.

[0019]FIG. 9 is an example of the configuration of the low pass filter 104. As shown in the figure, the transfer function of the filter is given by the formula below: $\begin{matrix} {\frac{Vlp}{Vi} = \frac{\left( {g_{m}/C} \right)}{s + \left( {g_{m}/C} \right)}} & (5) \end{matrix}$

[0020] The pole frequency ω₀ can be obtained as below. $\begin{matrix} {\omega_{0} = \frac{g_{m}}{C}} & (6) \end{matrix}$

[0021] Realization of an equalizing filter having a constant group delay characteristic and a variable cut-off frequency is attained by setting the pole frequencies ω₀ and Q of the biquadratic filters so as to satisfy the ratios of the pole frequencies and Q factors shown in FIG. 6. It is normally attained by controlling the g_(m) of the biquadratic filters. According to formulas (1) to (5), it is theoretically possible to make the cut-off frequencies of the filters variable by changing the capacitances C, however, integration of variable capacitors such as varicaps in a standard CMOS production process should be normally avoided since it increases the number of steps in the production process and leads to a rise of costs. Note that in this case as well, it is general practice to change the capacitances connected to the Gm-C integrators in stages of for example units of C such as C, 2C, and 3C or in units of 0.5C so as to coarsely change the cut-off frequency of a filter in a relatively wide range and then to control the g_(m) value to continuously and finely adjust the cut-off frequency.

[0022] When the capacitance C is fixed, the two values ω₀ and Q can be determined by adjusting the parameters of g_(m1), g_(m2), and g_(m3). Normally, the general practice is to set g_(m2)=g_(m3) or g_(m1)=g_(m3). By setting g_(m2)=g_(m3), the formulas (2) and (4) can be rewritten to the formulas below: $\begin{matrix} {{\omega_{0} = \frac{\sqrt{g_{m1}g_{m2}}}{C}},{Q = \sqrt{\frac{g_{m1}}{g_{m2}}}}} & (7) \end{matrix}$

[0023] On the other hand, when g_(m1)=g_(m3), the formulas (2) and (4) can be rewritten as follows: $\begin{matrix} {{\omega_{0} = \frac{g_{m1}}{C}},{Q = \frac{g_{m1}}{g_{m2}}}} & (8) \end{matrix}$

[0024] In each of the above cases, ω₀ can be also linearly changed while maintaining Q constant by linearly changing all g_(m)'s of the biquadratic filter. For example, if doubling both g_(m1) and g_(m2), the cut-off frequency ω₀ can be changed to be doubled while maintaining Q unchanged.

[0025] As will be understood from the above explanation, when designing a Gm-C configuration biquadratic filter and first-order low pass filter, it is preferable that the g_(m) values of the Gm-C integrators which compose whole filter can be controlled by an external means and can be controlled linearly.

[0026] In the related art, a silicon bipolar element has been used for such high frequency filter applications. In a bipolar element, as is well known, the g_(m) (hereinafter expressed by g_(m,bip) to distinguish it from the g_(m) of a MOS transistor) is indicated by the formula below: $\begin{matrix} {g_{m,\quad {bip}} = {{\frac{\partial\quad}{\partial V_{BE}}\left( {I_{s}\exp \quad \frac{V_{BE}}{V_{T}}} \right)} = {{\frac{I_{S}}{V_{T}}\exp \frac{V_{BE}}{V_{T}}} = \frac{I_{C}}{V_{T}}}}} & (9) \end{matrix}$

[0027] According to formula (9), the g_(m,bip) of a bipolar transistor is proportional to the collector current I_(C), so the g_(m,bip) can be linearly changed relatively easily.

[0028] On the other hand, a PRML read channel function is inseparable from the above Viterbi decoding or following digital processing as represented by the Reed Solomon error correction algorithm. There is a strong demand for analog PR equalization in a CMOS production process coexisting on the same die along with such purely digital processing blocks.

[0029] As is well known, the g_(m) of a MOS element (hereinafter expressed as g_(m,MOS) for clarification) is indicated by the formula below: $\begin{matrix} {g_{m,{MOS}} = {{\frac{\partial}{\partial V_{GS}}\left( {K\left( {V_{GS} - V_{th}} \right)}^{2} \right)} = {{2\sqrt{{KI}_{D}}} = {2{KV}_{eff}}}}} & (10) \end{matrix}$

[0030] Here, V_(th) is the threshold voltage of a MOS transistor, K=μC_(0X)/2L, and V_(eff)=V_(GS)−V_(th).

[0031] When comparing formula (10) and formula (9), the g_(m) of a bipolar element is linear with respect to the collector current I_(C), while the g_(m) of a CMOS element is linear with respect to the square root of the drain current I_(D). Thus, in the case of a CMOS element, when controlling the g_(m) by changing the drain current I_(D) by some external means, it is normally preferable in terms of controllability to provide a conversion mechanism for compensating for the above root characteristic for each of the transconductors and to change linearly from the minimum g_(m,min) to the maximum g_(m,max) in the variable range.

[0032] Summarizing the problems to be solved by the invention, in an equalizing filter configured by CMOS elements of the related art, for example, the cut-off frequency of the PR equalizing filter is adjusted by an adjusting means of about 5 to 6 bits width. In this case, a conversion mechanism for correcting the above root characteristic can also be realized by for example processing on a digital domain by a lookup table, and the root characteristic can be compensated for in an analog domain as well.

[0033] However, in any case, this leads to an increase of the circuit area due to the additional circuits and an increase of power consumption and causes deterioration of the filter characteristic itself.

[0034] Also, as another disadvantage, when controlling the g_(m) of a CMOS element by the drain current I_(D), some kind of linearizing means is generally required. Consequently, there arises a disadvantage that the inherent transconductance of a CMOS element cannot be obtained as a circuit g_(m). Furthermore, an increase of the circuit area and an increase of power consumption are inevitable.

[0035] For example, in a differential circuit comprised of MOS transistors shown in FIG. 10, the g_(m)'s of the MOS transistors can be controlled by a bias current I_(B). The relationship is expressed by the formula below: $\begin{matrix} {{{I1} - {I2}} = {{K\left( {{V1} - {V2}} \right)}\sqrt{\frac{2I_{B}}{K} - \left( {{V1} - {V2}} \right)^{2}}}} & (11) \end{matrix}$

[0036] As shown in formula (11), a non-linear term is included. As a result, linearization by a variety of linearizing means is necessary. The g_(m)'s obtained thereby becomes much smaller than the inherent g_(m,MOS) values of the MOS elements indicated in formula (10).

SUMMARY OF THE INVENTION

[0037] An object of the present invention is to provide a transconductor and a filter circuit capable of suppressing an increase of circuit area, enabling easy circuit design, realizing a reduction of power consumption by a common control voltage operation, and able to stably control the cut-off frequency.

[0038] To attain the above object, according to a first aspect of the present invention, there is provided a transconductor comprising a differential circuit including a first MOS transistor having a gate connected to a positive input terminal, a source grounded, and a drain connected to a first current source supplying a first operation current, and a second MOS transistor having a gate connected to a negative input terminal, a source grounded, and a drain connected to a second current source supplying a second operation current and including a control circuit for controlling the output currents of the first and second current sources in accordance with common-mode output voltage of the differential pair (which is comprised of the first and the second MOS transistors) and a predetermined reference potential.

[0039] According to a second aspect of the present invention, there is provided a filter circuit comprising a transconductor-C circuit (Gm-C circuit) including at least two transconductors and a load capacitor driven by the transconductors and a common-mode voltage control circuit for supplying a control signal for setting output common-mode voltage of the transconductors to be identical. Each transconductor includes a first MOS transistor having a gate connected to a positive input terminal, a source grounded, and a drain connected to a first current source supplying a first operation current and a second MOS transistor having a gate connected to a negative input terminal, a source grounded, and a drain connected to a second current source supplying a second operation current.

[0040] Preferably, the common-mode voltage control circuit generates the control signal in accordance with common-mode output voltage of the differential pair comprised of the first and second MOS transistors and a predetermined reference potential.

[0041] More preferably, the common-mode voltage control circuit generates the control signal so that an average voltage of the output signals output from the differential circuit comprised of the first and second MOS transistors becomes identical to the reference voltage.

[0042] Preferably, the common-mode voltage to be input to the gates of the first and second transistors is controlled to a desired transconductance value.

[0043] More preferably, the ratios of channel width and channel length of the first and second transistors are set to desired transconductance values.

BRIEF DESCRIPTION OF THE DRAWINGS

[0044] These and other objects and features of the present invention will become clearer from the following description of the preferred embodiments given with reference to the accompanying drawings, in which:

[0045]FIG. 1 is a circuit diagram of an embodiment of a filter circuit according to the present invention;

[0046]FIG. 2 is a circuit diagram of the configuration of a transconductance cell forming part of a filter circuit of the present invention;

[0047]FIG. 3 is a graph of a gain characteristic of a filter circuit of the present invention;

[0048]FIG. 4 is a graph of a group delay characteristic of a filter circuit of the present invention;

[0049]FIG. 5 is a block diagram of an example of the configuration of an equalizing filter;

[0050]FIG. 6 is a view of pole frequencies and Q factors of filter components;

[0051]FIG. 7 is a circuit diagram of the configuration of a biquadratic filter forming part of an equalizing filter;

[0052]FIG. 8 is a circuit diagram of the configuration of a biquadratic filter/equalizer forming part of an equalizing filter;

[0053]FIG. 9 is a circuit diagram of the configuration of a low pass filter forming part of an equalizing filter; and

[0054]FIG. 10 is a circuit diagram of an example of the configuration of a transconductance cell of the related art.

DESCRIPTION OF THE PREFERRED EMBODIMENTS

[0055] Below, preferred embodiments will be described with reference to the accompanying drawings.

[0056]FIG. 1 is a circuit diagram of an embodiment of a filter circuit according to the present invention.

[0057] As shown in the figure, a filter circuit of the present embodiment comprises transconductance cells (hereinafter referred to as g_(m) cells for convenience), capacitors, and bias circuits. Note that the filter circuit is for example a biquadratic filter used as an equalizing filter or a low pass filter.

[0058]FIG. 1 shows an embodiment of a biquadratic filter comprised of g_(m) cells. As shown in the figure, the filter circuit comprises g_(m) cells 10, 20, 30, and 40 and bias circuits 50 and 60.

[0059] The g_(m) cells 10, 20, 30, and 40 are transconductance cells each comprised of CMOS transistors and formed to have a g_(m) and Q designed in advance. The bias circuits 50 and 60 are comprised of common mode feedback circuits (CMFB) for supplying common-mode feedback voltages to the cells 10 and 20, and 30, and 40.

[0060] The capacitor C1 is driven by the g_(m) cells 10 and 20, while the capacitor C2 is driven by the g_(m) cells 30 and 40.

[0061] As shown in the figure, the g_(m) cell 20 and g_(m) cell 40 are serially connected. An input signal V_(i) is applied to an input side of the g_(m) cell 20. An output terminal of the g_(m) cell 20 is connected to the capacitor C1, while an output terminal of the g_(m) cell 40 is connected to a capacitor C2. Also, the g_(m) cell 10 and the g_(m) cell 20 are connected in parallel and an output signal V_(1p) of the g_(m) cell 40 is applied to an input terminal thereof. The g_(m) cell 30 and the g_(m) cell 40 are connected in parallel and the output signal V_(1p) of the g_(m) cell 40 is applied to its input side.

[0062] Namely, in the filter circuit shown in FIG. 1, the capacitor C1 is a load capacitance of the g_(m) cells 10 and 20, and the capacitor C2 is a load capacitance of the g_(m) cells 30 and 40.

[0063] Note that in the configuration shown in FIG. 1, the capacitors C1 and C2 forming load capacitances of g_(m) cells are expressed as differential capacitances, but when generally considering a parasitic capacitance, for example, if the load capacitance is “C”, it is preferable that the load capacitances be constituted by 2C capacitors connected between the positive and negative output terminals of the g_(m) cells and a ground potential for the reason of the ease of setting of the capacitance value or providing a function of phase compensation capacitance in a common-mode feedback loop etc.

[0064] The bias circuit 50 generates a bias voltage V_(bs1) in accordance with outputs V₀ of the g_(m) cell 20 and g_(m) cell 10 and a common-mode voltage setting terminal voltage (control voltage V_(c)) input from the outside and supplies the same to the g_(m) cells 10 and 20. Also, the bias circuit 60 generates a bias voltage V_(bs2) in accordance with outputs V_(1p) of the g_(m) cell 40 and g_(m) cell 30 and the control voltage V_(c) input from the outside and supplies the same to the g_(m) cells 30 and 40.

[0065] Note that the bias circuits 50 and 60 generate the bias voltage V_(bs1) and bias voltage V_(bs2) so that an average voltage of the common-mode components of output signals output from the g_(m) cells is identical to the control voltage V_(c).

[0066] The filter circuit having the configuration explained above utilizes the fact that the g_(m)'s of the CMOS elements change linearly with respect to V_(eff)=V_(GS)−V_(th), namely, makes the V_(GS)'s at an operation point of the MOS elements determining the g_(m)'s of the g_(m) cells a common potential, sets the K's of the MOS elements determining the g_(m) values, specifically, the channel widths W, in advance for every element so that the biquadratic filters or the low pass filter constituting the filter satisfy the pole frequencies and Q factors indicated in FIG. 6, and then lays out the elements. The cut-off frequency of the filter is adjusted by changing the common potential V_(GS).

[0067] The g_(m) cells constituting the filter circuit in FIG. 1 can be comprised by differential circuits comprised of MOS transistors as shown in FIG. 2.

[0068] As shown in FIG. 2, each g_(m) cell comprises MOS transistors M1 and M2 and current sources IS1 and IS2 for supplying an operation current to the transistors. Gates of the transistors M1 and M2 are connected to a positive input terminal and a negative input terminal of the g_(m) cell, while the sources are both grounded. A drain of the transistor M1 is connected to the current source IS1, while a drain of the transistor M2 is connected to the current source IS2. Output currents of the current sources IS1 and IS2 are controlled by a bias voltage V_(bs) supplied from the bias circuit 50 or 60 (for example, V_(bs1) or V_(bs2) shown in FIG. 1). As shown in FIG. 2, the amount of change i_(f) of the in-phase current component is supplied to the transistors M1 and M2 by the current sources.

[0069] In the illustrated g_(m) cell, the currents i₁′ and i₂′ are differential currents output by a differential pair of the transistors M1 and M2. Thus, an output differential current i_(od) of the g_(m) cell is given by the formula below: $\begin{matrix} \begin{matrix} {i_{od} = \quad {{i_{1}^{\prime}\left( {= {- i_{2}^{\prime}}} \right)} = {\frac{i_{1}^{\prime} - i_{2}^{\prime}}{2} = \frac{i_{1} - i_{2}}{2}}}} \\ {= \quad {{K\left( {V_{C} - V_{th}} \right)}\left( {v_{1} - v_{2}} \right)}} \end{matrix} & (12) \end{matrix}$

[0070] As will be understood from formula (12), the output differential current i_(od) of a g_(m) cell linearly changes with respect to the differential input voltage v₁−v₂.

[0071] As explained above, by constituting a biquadratic filter or a low pass filter by using the g_(m) cells shown in FIG. 2 and further by constituting the filter circuit (equalizing filter) of the present embodiment shown in FIG. 1, the cut-off frequency of the equalizing filter is controlled by a bias voltage supplied by the bias circuit connected in common to the g_(m) cells. Furthermore, since the g_(m) ratios of the g_(m) cells are kept constant and therefore the Q factors and group delay characteristics of the biquadratic filters and the low pass filter are maintained, the ratios of pole frequencies of the biquadratic filters can be kept constant. Since the g_(m)'s of the g_(m) cells linearly change with respect to a control voltage input from the outside, a function conversion mechanism is not necessary.

[0072] Furthermore, as shown in FIG. 2, by using a differential pair of source-grounded MOS transistors as a g_(m) cell, a g_(m) which is theoretically linear with respect to an input amplitude can be obtained.

[0073] Below, the operation of the filter circuit of the present embodiment comprised of the above g_(m) cells will be explained.

[0074] The g_(m) cells 10 and 20 driving the capacitance load C1 are supplied with a bias voltage V_(bs1) by the common bias circuit 50. Due to this, feedback control is performed so that common-mode output voltage of the g_(m) cells become desired values set by the bias voltage V_(bs1).

[0075] Also, similarly, the gm cells 30 and 40 driving the capacitance load C2 are supplied with a bias voltage Vbs2 by the common bias circuit 60. Due to this, feedback control is performed so that common-mode output voltage in the g_(m) cells become desired values set by the bias voltage V_(bs2).

[0076] Furthermore, as shown in FIG. 1, output terminals of the g_(m) cells are connected to other g_(m) cells or, in some cases, to their own input terminals. As a result, input/output terminals of all g_(m) cells operate at a common-mode voltage. The voltage is set in common by the bias voltage V_(bs1) or V_(bs2) given to the common-mode voltage setting terminals of the g_(m) cells.

[0077] Here, the transconductances g_(m,MOS) of a differential pair of MOS elements, for example, MOS transistors, are given by the above formula (10). According to the formula, it is understood that the g_(m) of a MOS element changes linearly with respect to a voltage V_(eff) equal to V_(GS)−V_(th), that is, a difference of a voltage V_(GS) between a gate and source of the MOS transistor and its threshold voltage V_(th). The present embodiment uses this characteristic, that is, uses the bias voltage commonly set by the bias circuits 50 and 60 to be applied between the gates and sources of the MOS transistors of the g_(m) cells and uses the source-grounded MOS transistors as input transistors of the g_(m) cell. As a result, a g_(m) which is linear with respect to a common-mode voltage, that is, the voltage V_(GS) between the gates and sources of the MOS transistors, can be obtained.

[0078] Here, the biquadratic filter 103 in FIG. 5 comprising the g_(m) cells shown in FIG. 2 will be explained. Note that the same explanation can be applied to other components in FIG. 5, for example, the biquadratic filter/equalizer 101, the biquadratic filter 102, and the low pass filter 104.

[0079] As shown in FIG. 6, in the biquadratic filter 103, the normalized pole frequency is 2.3174 Hz and the Q factor is 2.0229, that is, Q>1. Therefore, the cut-off frequency can be made smaller in the g_(m) cell for providing g_(m1) when g_(m3)=g_(m1) than when g_(m3)=g_(m2). This becomes generally advantageous in terms of power consumption. Here, an explanation will be made assuming g_(m3)=g_(m1).

[0080] At this time, since ω₀=g_(m1)/C and Q=g_(m1)/g_(m2), for example, when the cut-off frequency of the overall equalizing filter is set to be 100 MHZ and C=1 pF, g_(m1)=1.4553 mS and g_(m2)=719.4 μS. From formula (10), when the sizes of the MOS elements are set so as to satisfy K₁/K₂=1.4553/0.7194 by setting g_(m1)=2K₁ V_(eff) and g_(m2)=2K₂ V_(eff), the cut-off frequency f_(c) of the filter circuit can be adjusted by changing the control voltage V_(eff) commonly used by the g_(m) cells without changing the ratios of the g_(m) values, that is, the Q factors.

[0081] Note that it is necessary that the pole frequencies of the equalizing filters always maintain the ratios shown in FIG. 6, thus the g_(m) values of all of the g_(m) cells have to always maintain fixed ratios with each other. Therefore, in the filter circuit of the present embodiment, the K's of the MOS elements for determining the g_(m) values of the g_(m) cells, actually, the channel widths W of the MOS transistors, have to be set to the above fixed ratio. By doing so, it is possible to control the cut-off frequency of the overall equalizing filter just by changing the voltage V_(GS) at an operation point or the common-mode voltage commonly used by the g_(m) cells. Function conversion is not necessary.

[0082] Also, when determining the K's of the MOS transistors for determining the g_(m) values, the channel length L can be made a parameter. However, since there is a tendency toward shorter channels to obtain larger g_(m) values, the channel lengths L and the g_(m) values tend to deviate from the proportional relationship due to the short channel effect. Thus, the present invention is not limited to the above. The K's of the g_(m) cells can be determined by either of the channel lengths L or the channel widths W of the MOS elements and the g_(m) values can be determined thereby.

[0083]FIG. 3 is a graph of the gain characteristic of a filter circuit of the present embodiment and shows the change of the gain characteristic of the filter when changing a V_(c) terminal voltage.

[0084] Also, FIG. 4 is a graph of the group delay characteristic of the filter circuit of the present embodiment and shows the change of the group delay characteristic of the filter when changing a V_(c) terminal voltage.

[0085] Summarizing the effects of the present invention, as explained above, according to the filter circuit of the present invention, the transconductances (gm) of the transconductors of a filter are controlled by a common control voltage. Therefore, the cut-off frequency of the filter circuit can be easily adjusted and a function conversion mechanism using a lookup table, a function conversion circuit, etc. is totally unnecessary. As a result, it is possible to simultaneously achieve a reduction of circuit area, a reduction of power consumption, and prevention of additional disturbance which may be caused by additional circuits.

[0086] Also, in the present invention, the Q factors of the biquadratic filters comprised of the transconductors are kept unchanged since the ratios of gm values between the Gm-C integrators, which become important to achieve the group delay characteristic of the filter, are determined by the ratios of the shape parameters of the MOS elements, that is, the channel width W and channel length L, which are not affected by aging etc., so stable adjustment of the cut-off frequency can be attained.

[0087] Furthermore, according to the transconductor of the present invention, the output common-mode voltage becomes the same potential as an externally set common-mode voltage control voltage, and the output common-mode voltage becomes an input common-mode voltage of another transconductor in the next stage or connected to the output terminal thereof, so all transconductors constituting the overall filter operate at an identical common-mode voltage. As a result, there are advantages that an operation point of the overall filter circuit can be easily designed, a dynamic range of input and output can be enlarged, and a power consumption can be lowered.

[0088] Note that the present invention is not limited to the above embodiments and includes modifications within the scope of the claims. 

What is claimed is:
 1. A transconductor comprising: a differential circuit including a first MOS transistor having a gate connected to a positive input terminal, a source grounded, and a drain connected to a first current source supplying a first operation current, and a second MOS transistor having a gate connected to a negative input terminal, a source grounded, and a drain connected to a second current source supplying a second operation current and a control circuit for controlling the output currents of said first and second current sources in accordance with output signals output by said differential circuit and a predetermined reference potential.
 2. A transconductor as set forth in claim 1, wherein said control circuit controls the output currents of said first and second current sources so that the average potential of said output signals is identical to said reference potential.
 3. A transconductor as set forth in claim 1, wherein the common-mode potentials input to the gates of said first and second transistors constituting said differential circuit are controlled to be identical to a desired transconductance value.
 4. A transconductor as set forth in claim 1, wherein the ratios of channel width and channel length of said first and second transistors constituting said differential circuit are set to be identical to desired transconductance values.
 5. A filter circuit comprising: a transconductor-C circuit (Gm-C circuit) including at least two transconductors and a load capacitance element driven by said transconductors and a common-mode potential control circuit supplying control signals for setting the output common-mode potentials of said transconductors to be identical, said transconductors including a first MOS transistor having a gate connected to a positive input terminal, a source grounded, and a drain connected to a first current source supplying a first operation current and a second MOS transistor having a gate connected to a negative input terminal, a source grounded, and a drain connected to a second current source supplying a second operation current.
 6. A filter circuit as set forth in claim 5, wherein said common-mode potential control circuit generates said control signal in accordance with output signals output from the differential circuit comprised of said first and second MOS transistors and a predetermined reference potential.
 7. A filter circuit as set forth in claim 6, wherein said common-mode potential control circuit generates said control signal so that an average potential of output signals output from the differential circuit comprised of said first and second MOS transistors becomes identical to said reference potential.
 8. A filter circuit as set forth in claim 5, wherein the common-mode potential to be input to the gates of said first and second transistors is controlled to be a desired transconductance value.
 9. A filter circuit as set forth in claim 5, wherein the ratios of channel width and channel length of said first and second transistors are set to be desired transconductance values. 